Characteristics of Wiring in Homes and Buildings
In many instances, it is desirable to install communications networks in homes and businesses using the pre-existing wiring. Utilizing the pre-existing wiring allows the homeowner or business owner to network the building using the existing copper infrastructure without a major investment in the installation of optical fiber or other network transmission media. However, the network also needs to be capable of transmitting data at high data rates.
The pre-existing wiring (i.e., telephone wiring and power wiring) of most homes and other buildings is not of uniform type and may consist of 24 gauge twisted quad wiring, unshielded flat pair, or other miscellaneous types of wiring. This wiring can produce severely distorted transmission channels. FIG. 1 shows an example of a network 100 using existing 24 gauge twisted copper, such as the existing telephone lines in a home or business. Network 100 includes a main line 101 and trunk lines 102, 103 and 104, which are each coupled at one end to main line 101. Main line 101 includes a signal source 105 at one end and a receiver terminator 106 at the opposite end. Receiver terminator 106 provides main line 101 with a 100 Ohm termination. In FIG. 1, main line 101 is 360 feet long. Trunk line 102 is 80 feet long and is coupled to main line 101 at a point 170 feet from signal source 105. Trunk line 103 is 25 feet long and is coupled to main line 101 at a point 90 feet from receiver terminator 106. Trunk line 104 is 25 feet long and is coupled to main line 101 at a point 40 feet from receiver terminator 106. Trunk lines 103 and 104 each have open, un-terminated ends (i.e., infinite termination) opposite the end that is coupled to main line 101. Trunk line 102 includes a 100 ohm terminator at an end opposite the end of trunk line 102 that is coupled to main line 101. Other examples of networks can include any number of terminated, un-terminated or improperly terminated lines.
FIG. 2 shows the frequency response of the transmission channel between signal source 105 and receiver terminator 106 of network 100 shown in FIG. 1. The un-terminated trunk lines, trunk lines 103 and 104, cause a deep null in the spectrum of the frequency response. Other networks may have multiple spectral nulls or a differently shaped frequency response.
Other sources of spectral nulls or distortions in the frequency response of a transmission channel include filters to reject interference from HAM radio bands. FIG. 3 shows the combined response of transmit and receive filters in a passband modulated transceiver, including RFI suppression filters, for a transmission band of between 4 MHz and 10 MHz. The spectral null in the center of the spectrum suppresses the 40 meter HAM band.
As long as the signal-to-noise ratio of a received signal is sufficiently high, channel distortion can be corrected by equalization. Near-optimal throughput can be achieved by using a decision-feedback equalizer or equivalent structure. (See G. D. Forney, Jr., and M. V. Eyuboglu, Combined Equalization and Coding Using Precoding, IEEE COMM. MAG., vol. 29, no. 12, pp. 25-34, December 1991.) An ideal decision-feedback equalizer (DFE) or equivalent precoding structure, in combination with a fractionally-spaced feedforward equalizer (FSE), can correct the distortion from a transmission channel in an optimal manner, enabling the achievable throughput to approach the theoretical channel capacity arbitrarily closely with the use of sufficiently complex coding schemes (See J. M. Cioffi, et al., MMSE Decision-Feedback Equalizers and Coding—Part I: Equalization Results, IEEE TRANS COMM., vol. 43, no. 10, pp. 2582-2594, October 1995; J. M. Cioffi, et al., MMSE Decision-Feedback Equalizers and Coding—Part II: Coding Results, IEEE TRANS COMM., Vol. 43, no. 10, p. 2595-2604, October 1995).
However, when the transmission band of the channel contains deep spectral nulls and the signal-to-noise ratio is low, a large part of the transmission band may become unusable. This can easily happen when the transmitted signal power is limited and the spectrum of the transmitted signal is constrained within a narrow bandwidth to allow spectral compatibility with other signals on the transmission channel. In cases in which the power spectral density (PSD) is constrained or in which the SNR is limited by self-crosstalk, the frequency-dependent SNR is fixed and the SNR cannot be improved by increasing the transmit PSD. FIG. 4 shows a combined response of the transmit and receive filters of FIG. 3 and the transmission channel between signal source 105 and receiver termination 106 of FIG. 1. In FIG. 4, much of the spectrum is unusable because it is near or below the noise floor of −120 dBm/Hz.
In such cases where the signal-to-noise ratio is relatively low and the channel contains large spectral nulls, the achievable throughput for traditional single-carrier modulation using integral bits per symbol may be zero. For example, a single-carrier transceiver operating with a baud rate of 4 Mhz, a 15 dB gap (a measure of the difference between the theoretical channel capacity and the achievable channel capacity) and integer bits per symbol on a channel having the power spectral density shown in FIG. 4 has an achievable capacity of zero bits per symbol. Therefore, traditional single-carrier modulation fails. Single carrier modulation schemes are further discussed below. Theoretical capacity, achievable capacity, and the gap between them are further discussed below.
The problem of transmitting data through noisy channels having large spectral nulls is often solved by using transceivers that utilize either multicarrier modulation or frequency diverse modulation schemes. Multicarrier modulation or frequency diverse modulation schemes may provide acceptable throughput in such cases, but these schemes have additional implementation complexity and other practical disadvantages in comparison with single-carrier modulation transceivers.
Multi-Carrier Modulation
Multi-carrier modulation is a popular solution in some applications. The most common type of multi-carrier modulation is Discrete Multi-Tone (DMT) modulation. See J. A. Bingham, et al., Multicarrier Modulation for Data Transmission: An Idea Whose Time Has Come, IEEE COMM. MAG., May 1990, 5-14; I. Kalet, The Multitone Channel, IEEE TRANS. COMM., Vol. 37, No. 2, February 1989. On typical subscriber-loop channels, for example, DMT modulation generally achieves the same throughput as single-carrier modulation, assuming equivalent coding methods and properly optimized parameters. On severely distorted channels with large unusable spectral regions, however, DMT modulation transceivers may achieve better throughput than single-carrier modulation transceivers, especially when the capacity gap (see discussion of channel capacity below) is large.
DMT modulation transceivers have some disadvantages, however, as compared to single carrier modulation transceivers. A first disadvantage is that DMT modulation requires that the transmitter be informed of the transmission channel response. Therefore, DMT requires significant amounts of information flow from the receiver to the transmitter as well as data flow from the transmitter to the receiver. In addition, DMT modulation has a much higher peak-to-average ratio than single-carrier modulation, requiring the use of more expensive analog-to-digital and digital-to-analog converters with greater dynamic ranges than is required in single-carrier systems. DMT modulation also has less natural immunity to narrowband interference than single-carrier modulation. In addition, DMT modulation has a more complex transceiver structure compared to single-carrier modulation. These factors make DMT unattractive for many applications.
Frequency-Diversity
A frequency-diverse system is a system in which the transmitter of the transceiver modulates a signal with more than one carrier frequency, providing spectral redundancy in the transmitted signal (See T. RAPPAPORT, WIRELESS COMMUNICATIONS, PRINCIPLES AND PRACTICE, section 6.10.5 (1996)). The receiver of the frequency-diverse system then selects and demodulates the best frequency band or some combination of the different bands based on the characteristics of the transmission channel as measured at the receiver. Typically, the quality of the different frequency bands is unknown or time-varying because the network response is unknown or time-varying. A traditional frequency-diverse transceiver typically consists of two or more single-carrier transceivers in parallel. Although the transmitter does not require knowledge of the channel characteristics, the receiver must include additional logic to select the best frequency band. Additionally, the receiver requires a separate receiver structure for each modulation frequency, adding complexity to the receiver.
An example of a frequency-diverse QAM transceiver is shown in FIGS. 5A and 5B. FIG. 5A shows a transmitter 501 that transmits signals having multiple carrier frequencies ω1 through ωN. A host system 502 sends a symbol stream to transmitter 501. The symbol stream is split into its real and imaginary parts and filtered in transmit filter 506. Often, the input symbol stream is also upsampled and zerofilled in transmit filter 506. FIG. 6A shows an example of a power spectrum of a short sequence of data symbols transmitted at a baud rate of 1/T (i.e., T is the symbol period). In FIG. 6A, the solid line represents the base-band transmission spectrum and the dotted line is the repeated transmission spectrum that results from upsampling and inserting null samples between adjacent symbols. FIG. 6B shows the power spectrum of the complex output of a low-pass filter. In a QAM transceiver, for example, transmit filter 506 includes low-pass filters that yield the power spectrum shown in FIG. 6B given the signal power spectrum shown in FIG. 6A. In FIG. 6B, the spectral response of transmit filter 506 is shown by the dotted line. The spectral response of transmit filter 506 shown in FIG. 6B is of a 50% excess-bandwidth square-root raised cosine pulse.
In FIG. 5A, the real portion of the symbol stream is mixed with the functions cos(ω1t) through cos(ωNt) in mixers 508-1 through 508-N, respectively.
The imaginary part of the sample stream is mixed with the functions sin(ω1t) through sin(ωNt) in mixers 509-1 through 509-N, respectively. The output signals from mixers 508-1 through 508-N and 509-1 through 509-N are added in adder 510 and the sum is coupled to transmission channel 511. FIG. 6C shows the right-handed power spectrum of the real signal obtained by modulating the signal power spectrum shown in FIG. 6B by the carrier frequencies 1.0/T, 3.5/T and 6.0/T.
Transmitter 501, therefore, transmits each symbol of the symbol stream from host 502 onto transmission channel 511 N times using N different carrier frequencies. One or more frequency bands may be unusable, but in a well-designed system, it's unlikely that all bands would be unusable. The receiver can read the symbol from any of the N bands into which it is transmitted. Typically, the receiver chooses a particular band of transmission from which to receive signals based on an error analysis of the symbol stream received at the receiver.
FIG. 5B shows a receiver 512 for receiving the signals transmitted from transmitter 501 of FIG. 5A. Receiver 512 includes N individual receivers 513-1 through 513-N, one for each of the N modulation frequencies ω1 through ωN, respectively. The signal from transmission channel 511 is received into each of receivers 513-1 through 513-N. The signal is mixed with the function cos(ω1t) through cos(ωNt) in mixers 514-1 through 514-N, respectively, and filtered in receive filters 516-1 through 516-N, respectively. The signal from transmission channel 511 is also mixed with the function sin(ω1t) through sin(ωNt) in mixers 515-1 through 515-N, respectively, and filtered in receive filters 516-1 through 516-N, respectively. The output signals from receive filters 516-1 through 516-N combined into a real and imaginary portion and received by equalization/decider 517-1 through 517-N, respectively. The output signals from equalization/decider 517-1 through 517-N are received by receive host 518. Receiver host 518 receives the output signals from equalizer deciders 517-1 through 517-N and, based on a statistical analysis of the symbol stream, chooses a best symbol stream from the usable frequency bands. Other receiver hosts may take a weighted average of the output sample streams of equalizer deciders 517-1 through 517-N. The weighted average sample stream, then, is presented to a single equalizer structure that may include a decision feedback equalizer and a slicer.
The multi-tone solution and the frequency diversity solution, although capable of sending data through lossy transmission channels with large gaps in the available bandwidth, require the use of multiple modulators and demodulators. This adds complexity to the transceiver and increases cost, making it unattractive for some applications. There is a need for a single-carrier transceiver structure for transmitting signals through channels having large spectral nulls.
Characteristics of Wide-Band (WB) and Ultra-Wide-Band (UWB)
Ultra-wide-band refers to a type of wireless communication system that transmits energy over a very large bandwidth at a very low power-spectral density (PSD), relative to other non-UWB wireless systems. An 802.11a signal, which occupies about 20 MHz of bandwidth at a power spectral density (PSD) as high as 17 dBm/MHz, is an example of a typical non-UWB system. In contrast, under FCC rules, an ultra-wide-band signal must occupy at least 500 MHz of bandwidth, and it may occupy up to several GHz of bandwidth, at a maximum power spectral density of about −41.3 dBm/MHz. This difference in bandwidth and power spectral density naturally leads to modulation schemes that may be quite different from conventional narrowband wireless communications.
The ultra-wide-band environment also has several important differences when compared to the home phoneline environment. For example, the total available bandwidth for ultra-wide-band is much larger the home phoneline environment. This enables much larger ratios of signal bandwidth to symbol rate for ultra-wide-band. Additionally, the expected signal-to-noise ratio (SNR) is much lower, making higher-order quadrature amplitude modulation (QAM) constellations less useful. As a result, ultra-wide-band systems may typically employ quadrature phase shift keying (QPSK) or bi-orthogonal phase shift keying (BPSK), although other constellations may be utilized. Furthermore, an ultra-wide-band system is very likely to encounter interference from similar ultra-wide-band systems that may be operating in the same frequency band. Accordingly, while the frequency diverse single carrier modulation disclosed in U.S. Pat. No. 6,327,311 to Ojard is very useful for home phoneline environments, there is additional benefit to be gained by adapting it specifically to ultra-wide-band applications.
In a wireless personal area network (WPAN) environment, individual devices may form piconets. Within a single piconet, devices may communicate with each other, and communication may be coordinated among these devices so that the devices do not transmit at the same time. Although communication among devices within a piconet may be coordinated, communication among neighboring piconets may not be coordinated. Since neighboring piconets will not be coordinated, there is a strong potential for interference from adjacent uncoordinated piconets. Accordingly, it is highly desirable to minimize the effect of such interference.
Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of skill in the art, through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings.